Bidirectional high sidelobe suppression silicon optical phased array

Huaqing Qiu; Yong Liu; Xiansong Meng; Xiaowei Guan; Yunhong Ding; Hao Hu

doi:10.1364/PRJ.479880

Abstract

An optical phased array (OPA), the most promising non-mechanical beam steering technique, has great potential for solid-state light detection and ranging systems, holographic imaging, and free-space optical communications. A high quality beam with low sidelobes is crucial for long-distance free-space transmission and detection. However, most previously reported OPAs suffer from high sidelobe levels, and few efforts are devoted to reducing sidelobe levels in both azimuthal (

φ

) and polar (

θ

) directions. To solve this issue, we propose a Y-splitter-assisted cascaded coupling scheme to realize Gaussian power distribution in the azimuthal direction, which overcomes the bottleneck in the conventional cascaded coupling scheme and significantly increases the sidelobe suppression ratio (SLSR) in the

φ

direction from 20 to 66 dB in theory for a 120-channel OPA. Moreover, we designed an apodized grating emitter to realize Gaussian power distribution in the polar direction to increase the SLSR. Based on both designs, we experimentally demonstrated a 120-channel OPA with dual-Gaussian power distribution in both

φ

and

θ

directions. The SLSRs in

φ

and

θ

directions are measured to be

15.1 dB

and

25 dB

, respectively. Furthermore, we steer the beam to the maximum field of view of

25 ° \times 13.2 °

with a periodic

2 λ

pitch (3.1 μm). The maximum total power consumption is only 0.332 W with a thermo-optic efficiency of

2.7 mW / π

1. INTRODUCTION

An optical phased array (OPA), as a novel solid-state beam steering technique, is becoming an alternative to the mechanical beam steering method that has been used in commercial light detection and ranging (LiDAR) systems [1 –4]. OPAs are arrays of coherent optical emitters, with a working principle similar to the phased array antennas in radio waves. The far-field optical beam can be steered through the interference of emissions by controlling the phase of each emitter. Since OPA can be achieved on an integrated platform, it features a small size, weight, power, and cost (SWaP-C). It can be mass produced for various applications such as LiDAR [5 –8] and free-space optical communication [9]. To date, integrated OPAs have been achieved on various integrated platforms such as silicon (Si) [10 –17], Si nitride ( ${Si}_{3} N_{4}$ ) [18], lithium niobate ( ${LiNbO}_{3}$ ) [19], and indium phosphide (InP) [20]. Among them, Si is a desirable platform since it is fully compatible with the complementary metal oxide semiconductor (CMOS) fabrication process, allowing for integration with electronic controlling circuits [11,12].

Recently, research on integrated OPAs has mainly focused on scalability [11 –13,21], field of view (FoV) [13,14,22], high resolution [14], and low power consumption [23], while few works achieve a high sidelobe suppression ratio (SLSR). SLSR is a crucial parameter for integrated OPAs, which determines the beam quality and is a bottleneck for many applications such as long-distance free-space optical communication.

Conventional OPAs with uniform emission have a ${sinc}^{2}$ pattern in the far field, resulting in a theoretical minimum sidelobe level of $- 13.26 dB$ . By applying a Gaussian power distribution, sidelobes can be suppressed with the cost of a reduced effective emitting area. In 2019, Xie et al. demonstrated a 32-channel OPA and achieved a $\sim 16 - dB$ SLSR in azimuthal ( $φ$ ) direction using a star coupler for Gaussian amplitude distribution [24]. Moreover, there has been a recent demonstration that achieved 19 dB SLSR in azimuthal ( $φ$ ) direction for a 64-channel OPA by Liu and Hu using a star coupler as well [13]. However, to the best of our knowledge, applying Gaussian power distribution in both azimuthal ( $φ$ ) and polar ( $θ$ ) directions to achieve high sidelobe suppression has not been achieved yet.

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(a) Simulation results for uniform power distribution and Gaussian power distribution. (b) Simulated near field of uniform power distribution and Gaussian power distribution for φ direction, θ direction, and φ+θ direction.

Figure 1.(a) Simulation results for uniform power distribution and Gaussian power distribution. (b) Simulated near field of uniform power distribution and Gaussian power distribution for $φ$ direction, $θ$ direction, and $φ + θ$ direction.

Here, we propose novel Y-branch-assisted cascaded directional couplers and a specially designed apodized grating emitter to achieve Gaussian power distribution in both $φ$ and $θ$ directions simultaneously. We fabricate and demonstrate a 120-channel one-dimensional (1D) OPA with a periodic 3.1 μm pitch. A steering range of $25 ° \times 13.2 °$ ( $φ \times θ$ ) is achieved with the beam divergence of $0.31 ° \times 0.07 °$ . The total power consumption of the OPA is 0.332 W. The measured SLSRs in $φ$ and $θ$ directions are 15.1 and 25 dB, respectively.

2. DEVICE DESIGN

The formulas below describe the near-field and far-field patterns of the periodic OPA: $e (x) = g (x) \cdot \sum_{n = 1}^{N} δ (x - x_{n}),$ (1) $E (u) = F [g (x)] \otimes F [δ (x - x_{n})] = G (u) \otimes \sum_{n = 1}^{N} δ (u - n \frac{2 π}{d}),$ (2)where $x_{n}$ is the emitter position in x-space, and $e (x)$ and $E (u)$ describe the near-field and far-field patterns in x-space and u-space, respectively. The near field $e (x)$ of the periodic OPA can be regarded as the multiplication between the envelope $g (x)$ and the Dirac comb with a periodic pitch of $d$ . After Fourier transformation, the far-field pattern can be described by Eq. (2). Assuming that the power is equally distributed on each channel, the rectangular envelope of $g (x)$ in the near field will be Fourier transformed to the Sinc-function pattern in the far field. Therefore, the theoretical SLSR limitation is calculated to be 13.26 dB. An effective method to increase SLSR is by applying Gaussian power distribution in the near field. As a result, the far-field power distribution will be a Gaussian function as well, and the sidelobe can be suppressed. Figure 1(a) shows the simulated far-field power distribution for both uniform OPA (blue curve) and Gaussian OPA (red curve) with a periodic pitch of 3.1 μm. The SLSR is significantly increased to 25 dB when a 12 dB (center-to-edge ratio) Gaussian power distribution is applied to the near field instead of uniform power distribution. The simulated near-field power distributions are shown in Fig. 1(b). From the upper left to right bottom figure, the near field is changed from uniform to Gaussian power distribution in both directions, which can enable a high SLSR ( $25 dB \times 25 dB$ in theory).

A. $φ$ Direction: Y-Splitter-Assisted Cascaded Coupling

Figure 2.(a) Schematic of the through-type cascaded coupler with Gaussian power distribution. (b) Required power, residual power, and coupling efficiency for the largest Gaussian center-to-edge ratio. (c) The largest SLSR is only 20 dB for the through-type cascaded coupler. (d) Schematic of the Y-branch-assisted cascaded coupler with Gaussian power distribution. (e) Required power, residual power, and coupling efficiency for the largest Gaussian center-to-edge ratio. (f) The largest SLSR can achieve 66 dB for the Y-branch-assisted cascaded coupler.

An improved scheme is Y-junction-assisted cascaded couplers, as shown in Fig. 2(d). Different from through-type cascaded couplers, light is coupled from the center to both sides; therefore, the required power and residual power for the first coupler (i.e., No. 60 channel or No. 61 channel in the center) are the largest simultaneously. Consequently, the coupling efficiency is not limited by the minimum coupling efficiency. The calculated coupling efficiencies for different channels are shown in Fig. 2(e). The corresponding largest SLSR can reach 66 dB in theory. Furthermore, we explore the scalability of the two types of cascaded couplers: Y-junction-assisted cascaded couplers can support 1024 channels even with 12 dB 140 Gaussian power distribution (see Appendix A.3), while through-type cascaded couplers can support only 420 channels and only for uniform power distribution (see Appendix A.2).

Figure 3.(a) Designed 12 dB Gaussian power distribution on 120-channel OPA. (b) Far-field figure with SLSR of 25 dB. (c) Required power, residual power, and coupling efficiency for each channel. (d) Coupling length for each channel.

Figure 4.(a) Apodized grating emitter with Gaussian power distribution in the near field. (b) Zoomed-in figure in (a).

Even though Y-junction-assisted cascaded couplers can increase the SLSR up to 66 dB in theory by using 51.7 dB Gaussian power distribution, it is difficult to observe such a large SLSR using an infrared camera, which typically has a dynamic range of only 30 dB. In the experiment, we design 12 dB Gaussian power distribution [Fig. 3(a)] of the 120-channel OPA, which can theoretically realize a 25 dB SLSR in $φ$ direction [Fig. 3(b)]. The calculated coupling efficiency and coupling length for each channel are shown in Figs. 3(c) and 3(d), respectively.

B. $θ$ Direction: Apodized Grating Emitter

So far, few schemes have paid attention to Gaussian power distribution in $θ$ direction to increase the SLSR. One way to achieve this is an apodized grating emitter, which is usually used to decrease the coupling loss from the waveguide to the single-mode fiber (SMF) by reshaping the diffracted power distribution in the near field [25]. A schematic of the apodized grating emitter is shown in Fig. 4(a). We design the length, etching depth, and pitch $Λ$ of 2 mm, 10 nm, and 0.7 μm, respectively. The near-field power distribution can be reshaped by adjusting the ${SiO}_{2}$ duty cycle (etched part) for each emitting unit ( $Λ$ ), as shown in Fig. 4(b) (see Appendix A.4).

Figure 5.(a) Designed 12-dB Gaussian power distribution along 2-mm-long apodized grating emitter. (b) Corresponding SLSR in the far-field figure. (c) Simulated emitting efficiency per pitch and ${SiO}_{2}$ duty cycle along the apodized grating emitter. (d) The adjusted value of pitches guarantees the beam in the same direction.

Figure 6.(a) Schematic of proposed dual-Gaussian power distribution OPA. (b) Microscopy of proposed OPA. (c) SEM image of the apodized grating emitter with the largest ${SiO}_{2}$ duty cycle.

Considering the smallest feature size in the fabrication process, we design 12 dB Gaussian power distribution along the apodized grating emitter [Fig. 5(a)], which can achieve an SLSR of 25 dB [Fig. 5(b)]. The emitting efficiency of each unit and corresponding ${SiO}_{2}$ duty cycle are shown in Fig. 5(c). According to the grating formula, i.e., $n_{0} \sin θ_{0} + m \frac{λ}{Λ} = n_{eff}$ , the effective refractive index $n_{eff}$ will vary when the ${SiO}_{2}$ duty cycle changes. To precisely converge the far-field beam, the pitch of each unit along the apodized grating emitter is adjusted, and the result is shown in Fig. 5(d) (see Appendix A.4).

3. DEVICE FABRICATION

A schematic of the 120-channel OPA with Gaussian power distribution in both directions is shown in Fig. 6(a). The OPA consists of the coupler, Y-branch-assisted cascaded couplers, energy-efficient thermo-optic phase shifters [26,27], and apodized grating emitters. The optical path of each channel from the Y-branch to the apodized grating emitter is specially designed to be the same length to ease the phase alignment. Every three phase shifters are combined as a group and share a common ground to match the I/O ports of a commercial field programmable gate array (FPGA). The adjacent phase shifters are periodically placed with a minimum pitch of 107 μm to eliminate thermal cross talk.

The 120-channel OPA is fabricated on a commercial Si-on-insulator (SOI) chip. First, we utilize E-beam lithography (EBL) and deep reactive ion etching (DRIE) to pattern the fully etched passive waveguide. Second, we repeat the processes to pattern and etch the 10 nm deep apodized grating emitter. Third, a 1 μm thick layer of Si dioxide ( ${SiO}_{2}$ ) is deposited by plasma-enhanced chemical vapor deposition (PECVD) to clad the Si waveguide. The ${SiO}_{2}$ layer aims to protect the Si waveguide from being damaged, and decrease the optical loss caused by the metal layer that will be deposited on top. Fourth, we deposit the Ti layer as the micro-heater. Due to the large heater width (3 μm) designed in the phase shifter [27], ultraviolet (UV) lithography (375 nm wavelength) is used to define the heater pattern, followed by a layer of 110 nm thick Ti deposition using the E-beam evaporator (EBE) process. After the lift-off process, the Ti heater is fabricated. Last, we deposit a 500 nm thick layer of Au with the same process to act as the gold line to connect the chip with the printed circuit board (PCB). The microscope image is shown in Fig. 6(b). The average resistance is measured to be $\sim 536 Ω$ . Figure 6(c) indicates the scanning electron microscope (SEM) image of the apodized grating emitter. The largest etching width of the apodized grating emitter is 235 nm, which matches well with the designed value, i.e., $0.7 μm \times 0.3$ (largest ${SiO}_{2}$ duty cycle) = 210 nm.

4. EXPERIMENT

A schematic of the far-field measurement setup is shown in Fig. 7(a). The chip-PCB is mounted on the fixed stage, which is placed in the center of a round rotation stage. The infrared camera is mounted on the round rotation stage, which can achieve a flexible rotation of 360°. A convex lens is mounted in front of the infrared camera to realize Fourier transformation between the near-field and far-field figures. Therefore, the far-field figure can be obtained at a short distance, i.e., in the image focal plane of the convex lens. When the beam is steered in $φ$ direction, the camera can be rotated to the target angle to capture the emitted beam. Figure 7(b) illustrates the fabricated $2.1 cm \times 2.1 cm$ OPA chip, which is wire-bonded on a PCB. The input/output (I/O) ports on the PCB are connected to a commercial FPGA using jumper cables. Each cable has 26 electric lines and can produce 26 electric signals (signal and ground) simultaneously. Both FPGA and the infrared camera are connected to a computer via universal serial bus (USB) cables. In the experiment, the captured far-field image by the infrared camera was transferred to the computer. By analyzing the images, the computer will control the FPGA and change its output electric signals with the pulse-width-modulation (PWM) technique (see Appendix B.1), which can control the 120 phase shifters on the chip.

Figure 7.(a) Schematic of the measurement setup. (b) The $2.1 cm \times 2.1 cm$ silicon OPA chip is wire-bonded on a PCB.

With a wavelength of 1550 nm, the far-field image before calibration is shown in Fig. 8(a). Due to the design of an equal optical path for each channel, there exists a converged bright dot in the far field even though the phase shifters are not calibrated. Following the gradient descent algorithm, the brightest dot is chosen and optimized (see Appendix B.2). After calibration, the power of the beam will be more concentrated. In $φ$ direction, the far-field power distribution is shown in Fig. 8(b), which is obtained by collecting the maximum power on each $φ$ value. The calculated FoV is 29°, shown as the light-green area in Fig. 8(b). The measured SLSR and beam width in $φ$ direction are 15.1 dB and 0.31°, respectively. The measured far field in $θ$ direction is shown in Fig. 8(c). The measured SLSR and beam width in $θ$ direction are 25 dB and 0.07°, respectively, which match well with 25 dB and 0.05° in the simulation.

Figure 8.(a) Far-field figure before and after calibration. (b) Far-field figure in $φ$ direction; $φ$ SLSR is 15.1 dB at $φ = - 2.4 °$ . (c) Cross- section of the calibrated far-field figure; $θ$ SLSR is 25 dB.

After the far-field beam is calibrated, we vary the phase from 0 to $2 π$ for each channel and measure the intensity changes between the peak and the bottom, as shown in Fig. 9(a). The peak-to-bottom intensity change is proportional to the amplitude distribution for each channel [13]. We fit the intensity change with the Gaussian function and get 6 dB Gaussian amplitude distribution, which matches well with the design of 12 dB Gaussian power distribution. We then re-simulate the far-field figure with the measured Gaussian amplitude distribution, shown in Fig. 9(b). The simulated SLSR and beam width in $φ$ direction are 17.1 dB and 0.27°, respectively, which match well with the experimental results, i.e., 15.1 dB and 0.31°, respectively. Moreover, the degradation between theoretical (25 dB) and experimental (15.1 dB) values of the SLSR in $φ$ direction is mainly attributed to the imprecise power control using the cascaded coupler, the signal cross talk in the FPGA, noise of the PWM signal, and signal cross talk due to the common ground. The SLSR could be improved by replacing the FPGA with a digital to analog converter (DAC)-controlling system (see Appendices C.1 and C.2).

Figure 9.(a) Peak-to-valley far-field intensity vibration when phase varies from 0 to $2 π$ for each channel. (b) Simulated far-field figure with measured amplitude distribution of (a).

Furthermore, we tune the phase shifters to achieve beam steering in $φ$ direction, as shown in Fig. 10(a). The beam is steered from $- 11 °$ to 14°. The far-field power in $φ$ direction is shown in Fig. 10(b). The measured SLSR and beam width of each steered angle are shown in Fig. 10(c). The measured SLSR ranges from 12.3 to 15.1 dB, and the average beam width is 0.31°. Due to the element factor of an OPA, the SLSR of the main lobe at 0° is larger than that at other angular positions [13].

Figure 10.(a) Far-field figure of the beam steered in $φ$ direction. (b) Cross section in $φ$ direction; the maximum far-field intensity on each $φ$ value is obtained, and therefore, the largest sidelobe value is obtained. (c) Calculated SLSR and beam width when the beam is steered in $φ$ direction.

Beam steering in $θ$ direction is achieved by wavelength tuning, and the far-field image is shown in Fig. 11(a); $θ$ is 8° when the wavelength is 1550 nm, and $θ$ is steered from 16.7° to 3.5° when the wavelength is increased from 1496 to 1580 nm. The tuning range is 13.2° when the wavelength is tuned by 84 nm. The far-field power in $θ$ direction is shown in Fig. 11(b). The measured SLSR and beam width of each steered angle are shown in Fig. 11(c). The measured SLSR ranges from 18 to 31 dB, and the average beam width is 0.07° (the theoretical beam width for a 12 dB Gaussian power distribution apodized grating emitter with 2 mm is 0.05°). Finally, the beam is steered in both $φ$ and $θ$ directions simultaneously. The number “7” is formed in the far field by tuning both the wavelength and phase shifters simultaneously, as shown in Fig. 11(d). Moreover, we calculate the total power consumption of the 120-channel OPA to be 0.332 W with average thermo-optical efficiency of $2.7 mW / π$ for each channel (see Appendix B.3).

Figure 11.(a) Far-field figure of the beam steered in $θ$ direction when the input wavelength is changed from 1496 to 1580 nm. (b) Cross section in $θ$ direction. (c) Calculated SLSR and beam width when the beam is steered in $θ$ direction. (d) The beam is steered in both $φ$ and $θ$ directions simultaneously.

5. DISCUSSION

A high SLSR is essential for long-range lidar applications. Although Gaussian power distribution enlarges the beam width, it can effectively increase the SLSR. By designing 25 dB Gaussian power distribution, the beam width is only slightly increased, as shown in Fig. 1(a). Moreover, the beam width is inversely related to the aperture size ( $number of channels \times pitch$ ); therefore, the beam width can be reduced by increasing the aperture. In Table 1, we compare state-of-the-art OPA schemes based on uniform power distribution. This work shows a high SLSR in both directions, and the beam width has negligible broadening due to Gaussian power distribution since the product of beam width and aperture is similar to that of the OPA based on uniform power distribution [11,12,23,28].Table 1.

Performance Comparison among State-of-the-Art Periodic 1D OPAs

Year	2018 [11]	2020 [12]	2020 [23]	2021 [28]	2022^a [13]	2022^a [29]	2023 This Work
Platform	Si	Si	Si	${Si}_{3} N_{4}$	Si	Si	Si
Wavelength (nm)	1550	1550	1570	1550	1550	1550	1550
Number of channels	1024	8192	512	64	64	1000	120
Pitch (μm)	2	1	0.52	2.5	0.775	0.775	3.1
FoV ( $φ$ )	40°	100°	70°	35.5°	140°	160°	25°
SLSR (dB)	9	10	9	5	19	12	$15.1 \times 25$
Aperture (mm)	2	8	2	0.16	0.05	0.775	0.363
Beam width ( $φ$ )	0.03°	0.01°	0.15°	0.69°	2°	0.25°	0.31°

Gaussian power distribution realized by star coupler.

6. CONCLUSION

In this paper, we demonstrated a 120-channel OPA with 12 dB Gaussian power distribution in the near field for both $φ$ and $θ$ directions using novel Y-branch-assisted cascaded couplers and an apodized grating emitter, respectively. We experimentally measured the SLSR in $φ$ and $θ$ directions as 15.1 and 25 dB, respectively. A steering range of $25 ° \times 13.2 °$ with averaged beam widths of 0.31° and 0.07° was achieved with a total power consumption of 0.332 W.

Acknowledgment

Acknowledgment. The authors thank DTU Nanolab for support of the fabrication facilities and technologies.

APPENDIX A: NUMERICAL STRUCTURE DESIGN

1.Cascaded Coupler

n

Figure 12.(a) Schematic of cascaded couplers. (b) Simulated relationship between coupling efficiency and coupling length. (c) Experimental measurement results for designed 15 dB Gaussian power distribution along 64 channels.

Figure 13.Number of channel limitations for through-type cascaded couplers. (a) The maximum number of channels is 420 for uniform power distribution. (b) Corresponding coupling lengths. (c) Power distribution, residual power before the $n$ th channel, and coupling efficiency of the $n$ th channel curves. (d) Far-field figure of uniform power distribution in theory, whose SLSR is 13.26 dB.

Figure 14.Number of channel limitations for Y-splitter-assisted cascaded couplers. (a) The maximum number of channels can reach 1024 with 12 dB Gaussian power distribution. (b) Corresponding far-field pattern with SLSR of 25 dB. (c) Power distribution, residual power before the $n$ th channel, and coupling efficiency. (d) Coupling length distributions.

Figure 15.(a) Periodic grating emitter with a fixed ${SiO}_{2}$ duty cycle of 0.5. (b) Far-field figure of the grating with $Λ = 0.7 μm$ ; the emitting angle is $θ_{0} = 8.4 °$ . (c) Emitting efficiency per pitch unit versus ${SiO}_{2}$ duty cycle. (d) Grating $n_{eff}$ versus ${SiO}_{2}$ duty cycle. (e) Far-field emitting angle versus ${SiO}_{2}$ duty cycle.

n_{eff}

APPENDIX B: EXPERIMENT

1.Pulse-Width-Modulation Technique

The direct current (DC) output voltage of FPGA is 3.3 V and cannot be tuned. However, the voltage added to the phase shifter should be tuned to align the phase. We utilize the PWM technique to achieve voltage varying from 0 to 3.3 V. The principle is shown in Fig. 16 . A clock signal (230 MHz in the experiment) is utilized in FPGA. Moreover, certain values of the period-set and the wave-set are fixed. At the rising edge trigger of the clock signal, the value in the register called “counter” will be incremented by one. If the wave-set value is larger than the counter value, the PWM output will be one; otherwise, the output will be zero. Furthermore, if the increasing counter value is equal to the period-set, the counter value will be reset as zero, and a new period occurs next. Normally the period-set value is fixed, and the duty cycle of the PWM signal can be tuned by changing the wave-set value.

Figure 16.Principle of PWM technique.

Figure 17.Gradient algorithm optimization process.

Figure 18.(a) The far-field intensity changes when electric power increases from 0 to 7 mW, indicating power consumption of $2.7 mW / π$ . (b) Electric power consumption on each channel when the beam is steered 14° in $φ$ direction; the total power consumption is 332 mW for the proposed 120-channel OPA.

2.Gradient Algorithm Optimization

2.2 π

3.Total Power Consumption

2 π

APPENDIX C: NOISE ANALYSIS

1.PWM Signal Quality

φ

Figure 19.(a) Measured PWM signal [30]. (b) Every three I/O ports share a common ground in the designed OPA. (c) Improved electric circuits of individual ground.

2.Noise Level of Common Ground

536 Ω

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